- Research Article
- Open Access
LTE Adaptation for Mobile Broadband Satellite Networks
© Francesco Bastia et al. 2009
- Received: 31 January 2009
- Accepted: 30 July 2009
- Published: 5 November 2009
One of the key factors for the successful deployment of mobile satellite systems in 4G networks is the maximization of the technology commonalities with the terrestrial systems. An effective way of achieving this objective consists in considering the terrestrial radio interface as the baseline for the satellite radio interface. Since the 3GPP Long Term Evolution (LTE) standard will be one of the main players in the 4G scenario, along with other emerging technologies, such as mobile WiMAX; this paper analyzes the possible applicability of the 3GPP LTE interface to satellite transmission, presenting several enabling techniques for this adaptation. In particular, we propose the introduction of an inter-TTI interleaving technique that exploits the existing H-ARQ facilities provided by the LTE physical layer, the use of PAPR reduction techniques to increase the resilience of the OFDM waveform to non linear distortion, and the design of the sequences for Random Access, taking into account the requirements deriving from the large round trip times. The outcomes of this analysis show that, with the required proposed enablers, it is possible to reuse the existing terrestrial air interface to transmit over the satellite link.
- Long Term Evolution
- Forward Error Correction
- OFDM Symbol
- Channel Quality Indicator
- PAPR Reduction
Integrated terrestrial and satellite communication system is a paradigm that has been addressed for many years and that is at the fore front of the research and development activity within the satellite community. The recent development of the DVB-SH standard  for mobile broadcasting demonstrates that virtuous synergies can be introduced when terrestrial networks are complemented with a satellite component able to extend their service and coverage capabilities. A key aspect for the successful integration of the satellite and terrestrial components is the maximization of technological commonalities aimed at the exploitation of the economy of scale that derives from the vast market basis achievable by the integrated system. In order to replicate in 4G networks the success of the integrated mobile broadcasting systems, many initiatives are being carried out [2, 3] for the design of a satellite air interface that maximizes the commonalities with the 4G terrestrial air interface. These initiatives aim at introducing only those modifications that are strictly needed to deal with the satellite channel peculiarities, such, for example, nonlinear distortion introduced by the on-board power amplifiers, long round-trip propagation times, and reduced time diversity, while keeping everything else untouched. Specifically, it is important to highlight the different mobile channel propagation models between terrestrial and satellite environments. In fact, in terrestrial deployments, channel fades are typically both time and frequency selective, and are counteracted by the use of opportunistic scheduling solutions, which select for each user the time slots and the frequency bands where good channel conditions are experienced. On the other hand, satellite links are characterized by large round trip delay, which hinders the timeliness of the channel quality indicators and sounding signals, continuously exchanged between users and terrestrial base stations. Further, satellite channel fades are typically frequency-flat, due to the almost Line-of-Sight (LOS) nature of propagation in open area environments, thus alternative solutions have to be designed in order to increase the satellite link reliability.
In this framework, this paper investigates the adaptability of the 3GPP Long Term Evolution (LTE) standard  to the satellite scenarios. The 3GPP LTE standard is in fact gaining momentum and it is easily predictable to be one of the main players in the 4G scenario, along with other emerging technologies such as mobile WiMAX . Thanks to this analysis, we propose the introduction of few technology enablers that allow the LTE air interface to be used on a satellite channel. In particular, we propose the following:
(i)an inter-TTI (Transmission Time Interval) interleaving technique that is able to break the channel correlation in slowly varying channels by exploiting the existing H-ARQ facilities provided by the LTE physical layer;
(ii)the introduction of PAPR reduction techniques to increase the resilience of the OFDM waveform to nonlinear distortions;
(iii)a specific design of the sequences for the random access scheme, taking into account the requirements deriving from large satellite round trip times.
In addition, with the aim of further enhancing the robustness to long channel fades, an Upper-Layer (UL) Forward Error Correction (FEC) technique is also proposed and compared with the inter-TTI technique.
According to market and business analysis , two application scenarios are considered: mobile broadcasting using linguistic beams with national coverage and two-way communications using multispot coverage with frequency reuse. Clearly, the service typologies paired with these two application scenarios have different requirements in terms of data rates, tolerable latency, and QoS. This has been taken into account into the air interface analysis.
The 3GPP LTE air interface is shortly summarized to ensure self-containment and to provide the perspective for the introduction of advanced solutions for the adaptation to satellite links, as described in Section 3.
(1)Each of the three streams is interleaved separately by a sub-block interleaver.
(2)The interleaved systematic bits are written into the buffer in sequence, with the first bit of the interleaved systematic bit stream at the beginning of the buffer.
(3)The interleaved P1 and P2 streams are interlaced bit by bit. The interleaved and interlaced parity bit streams are written into the buffer in sequence, with the first bit of the stream next to the last bit of the interleaved systematic bit stream.
(4)Eight different Redundancy Versions (RVs) are defined, each of which specifies a starting bit index in the buffer. The transmitter reads a block of coded bits from the buffer, starting from the bit index specified by a chosen RV. For a desired code rate of operation, the number of coded bits to be selected for transmission is calculated and passed to the RM block as an input. If the end of the buffer is reached and more coded bits are needed for transmission, the transmitter wraps around and continues at the beginning of the buffer, hence the term of "circular buffer.'' Therefore, puncturing, and repetition can be achieved using a single method.
The CB has an advantage in flexibility (in code rates achieved) and also granularity (in stream sizes). In LTE, the encoded and interleaved bits after the RM block are mapped into OFDM symbols. The time unit for arranging the rate matched bits is the Transmission Time Interval (TTI).
Throughout all LTE specifications, the size of various fields in the time domain is expressed as a number of time units, seconds. Both downlink and uplink transmissions are organized into radio frames with duration ms. In the following, the Type-1 frame structure, applicable to both FDD and TDD interface, is considered. Each radio frame consists of 20 slots of length ms, numbered from 0 to 19. A sub-frame is defined as two consecutive slots, where sub-frame consists of slots and . A TTI corresponds to one sub-frame.
In general, the baseband signal representing a downlink physical channel is built through the following steps:
(i)scrambling of coded bits in each of the code words to be transmitted on a physical channel;
(ii)modulation of scrambled bits to generate complex-valued modulation symbols;
(iii)mapping of the complex-valued modulation symbols onto one or several transmission layers;
(iv)pre-coding of the complex-valued modulation symbols on each layer for transmission on the antenna ports;
(v)mapping of complex-valued modulation symbols for each antenna port to resource elements;
(vi)generation of complex-valued time-domain OFDM signal for each antenna port.
In the following sections, we propose and analyze some solutions to adapt the 3GPP LTE air interface to broadband satellite networks. These advanced techniques are applied to the transmitter or receiver side in order to enhance and maximize the system capacity in a mobile satellite environment.
3.1. Inter-TTI Interleaving
In this section, we propose an inter-TTI interleaving technique allowing to break channel correlation in slowly varying channels, achieved through the reuse of existing H-ARQ facilities provided by the physical layer of the LTE standard .
The LTE standard does not foresee time interleaving techniques outside a TTI . Thus, since the physical layer codeword is mapped into one TTI, the maximum time diversity exploitable by the Turbo decoder is limited to one TTI ( ). For low to medium terminal speeds, the channel coherence time is larger than , thus fading events cannot be counteracted by physical layer channel coding. In order to cope with such a fading events, LTE exploits both "intelligent" scheduling algorithms based on the knowledge of channel coefficients both in the time and in the frequency dimension, and H-ARQ techniques. The former technique consists in exploiting the channel state information (CSI) in order to map data into sub-carriers characterized by high signal to noise ratio (good channel quality). Of course this technique shows great benefits when frequency diversity is present within the active subcarriers.
H-ARQ consists in the "cooperation" between FEC and ARQ protocols. In LTE, H-ARQ operation is performed by exploiting the virtual circular buffer described in Section 2. Orthogonal retransmissions can be obtained by setting the RV number in each retransmission, thus transmitting different patterns of bits within the same circular buffer. Of course, H-ARQ technique yields to great performance improvement when time correlation is present because retransmission can have a time separation greater than channel coherence time.
Unfortunately, neither of the aforementioned techniques can be directly applied to the satellite case due to the exceedingly large transmission delays, affecting both the reliability of the channel quality indicators and of the acknowledgements. Nevertheless, it is possible to devise a way to exploit the existing H-ARQ facilities adapting them to the satellite use. To this aim, we propose a novel forced retransmission technique, which basically consists in transmitting the bits carried in the same circular buffer within several TTIs, that acts as an inter-TTI interleaving. To do this, we can exploit the same mechanism as provided by the LTE technical specifications for the H-ARQ operations with circular buffer. For the explanation of this solution, the block diagram depicted in Figure 1 can be taken as reference. In this example, 4 retransmissions are obtained by using 4 different RVs, starting from 0 up to 3. Each of the 4 transmission bursts is mapped into different TTIs, spaced by . is a key parameter because it determines the interleaving depth and it should be set according to channel conditions and latency requirements.
It is straightforward to derive the maximum time diversity achievable by adopting such as technique. Let be the number of retransmissions needed to complete the transmission of a single circular buffer, the number of OFDM symbols transmitted in each retransmission, and the duration of OFDM symbols. (The duration of the OFDM symbol is intended to be the sum of the useful symbol and cyclic prefix duration.) We have that a codeword is spread over total protection time . Given the fact that the standard facilities are used, no additional complexity is introduced. The drawback involved with the use of such technique is the data rate reduction, brought about by the fact that one codeword is not transmitted in but in . A possible way to maintain the original data rate is to introduce in the terminals the capability of storing larger quantities of data, equivalent to the possibility to support multiple H-ARQ processes in terminals designed for terrestrial use. In this way, capacity and memory occupation grow linearly with the number of supported equivalent H-ARQ processes, and is upper bounded by the data rate of the original link without inter-TTI.
3.2. PAPR Reduction Techniques
As an example of using this simple approximation, which becomes increasingly tight increasing the FFT size, it is easy to check that a PAPR of 9 dB is exceeded with a probability of 0.5 assuming , while a PAPR of 12 dB is exceeded with a probability of .
This argument motivates the use of a PAPR reduction technique, in order to lower the PAPR and drive the satellite amplifier with a lower back-off. Power efficiency is at a prime in satellite communications, and an eventual reduction of the back-off implies an improvement in the link budget and an eventual increase of the coverage area. Amongst all requisites for PAPR reduction techniques (see [9, 10] for a general overview), the compatibility with the LTE standard is still fundamental. Secondly, the receiver complexity must not be significantly increased. Furthermore, no degradation in BER will be tolerated, because it would require an increased power margin. Finally, the PAPR reduction method will cope with the severe distortion given by the satellite: even if the amplifier has an ideal pre-distortion apparatus on-board, it is operated near to its saturation, where a predistorter could not invert the flat HPA characteristic. The cascade of an ideal predistorter and the HPA is the so-called ideal clipping or soft limiter. In such a scenario, if the PAPR is lower than the IBO the signal will not be distorted, while if the PAPR is significantly higher the signal will be impaired by non-linear distortion. Thus, the PAPR reduction technique should offer a good PAPR decrease for almost all OFDM symbols, rather than a decrease which can be experienced with a very low probability.
Several techniques have been proposed in the literature, and even focusing on techniques which do not decrease the spectral efficiency, the adaptation to satellite scenario remains an issue: this is the case of Tone Reservation [11–13], the intermodulation products of satellite amplifier prevent using this technique, while it is very popular in the wired scenario and when the amplifier is closer to its linear region. The Selected Mapping technique [14, 15], although easy and elegant, needs a side information at the receiver. The side information can be avoided, at expense of a significant computational complexity increase at the receiver. Companding techniques (see  and references therein) offer a dramatic reduction in PAPR and do not require complex processing. On the other hand, there is a noise enhancement, which turns out to be an important source of degradation at the very low SNRs used in satellite communications.
The Active Constellation Extension (ACE) technique  fulfills those requirements, moreover the power increase due to PAPR reduction is exploited efficiently, obtaining an additional margin against noise. The ACE approach is based on the possibility to dynamically extend the position of some constellation points in order to reduce the peaks of the time domain signal (due to a constructive sum of a subset of the frequency domain data) without increasing Error Rate: the points are distanced from the borders of their Voronoi regions. The extension is performed iteratively, according to the following procedure.
(1)Start with the frequency domain representation of a OFDM symbol.
(2)Convert into the time-domain signal, and clip all samples exceeding a given magnitude . If no sample is clipped, then exit.
(3)Reconvert into the frequency domain representation and restore all constellation points which have been moved towards the borders of their Voronoi regions.
(4)Go back to 2 until a fixed number of iteration is reached.
This algorithm is applied to data carriers only, excluding thus pilots, preamble/signalling and guard bands. In the performance evaluation of the algorithm, the amplitude clipping value is expressed in term of the corresponding PAPR, which is called PAPR-Target in the following.
The most critical point of this method is the choice of the clipping level : a large value for (which corresponds to an high PAPR-Target) will yield a negligible power increase and a poor convergence, since signal is unlikely to be clipped. On the opposite extreme, a very low clipping level will yield again a poor convergence and a negligible power increase. In fact, considering the above algorithm, almost all points will be moved by clipping in step-2 and then restored by the constellation constraint enforcing in step-3. A compromise value, which will lead to a PAPR around 5 or 6 dB is advisable, yielding a good convergence and a slight energy increase, due to the effectiveness of the extension procedure. Although there are other ACE strategies , the solution presented here is attractive because it can be easily implemented both in hardware and software, as reported in .
3.3. Random Access Signal Detection
The Random Access Channel (RACH) is a contention-based channel for initial uplink transmission, that is, from mobile user to base station. While the Physical RACH (PRACH) procedures as defined in the 3G systems are mainly used to register the terminal after power-on to the network, in 4G networks, PRACH is in charge of dealing with new purposes and constraints. In an OFDM based system, in fact, orthogonal messages have to be sent, thus the major challenge in such a system is to maintain uplink orthogonality among users. Hence both frequency and time synchronization of the transmitted signals from the users are needed. A downlink broadcast signal can be sent to the users in order to allow a preliminary timing and frequency estimation by the mobile users, and, accordingly a timing and frequency adjustment in the return link. The remaining frequency misalignment is due to Doppler effects and cannot be estimated nor compensated. On the other hand, the fine timing estimation has to be performed by the base station when the signals coming from users are detected. Thus, the main goal of PRACH is to obtain fine time synchronization by informing the mobile users how to compensate for the round trip delay. After a successful random access procedure, in fact, the base station and the mobile user should be synchronized within a fraction of the uplink cyclic prefix. In this way, the subsequent uplink signals could be correctly decoded and would not interfere with other users connected to the network.
ZC allocation for GEO satellite scenario.
Cell Radius [km]
Number of root ZC sequences
Number of cyclic shift per root sequence
150 (Near polar arctic circle)
300 (Near polar arctic circle)
500 (Near polar arctic circle)
In this section, we propose a UL-FEC technique working on top of the PHY layer. It is well known that channel coding can be performed at different layers of the protocol stack. Two are the main differences which arise when physical layer or upper layer coding is addressed: the symbols composing each codeword, and the channel affecting the transmitted codeword. Indeed, at physical layer the symbols involved in the coding process typically belong to the Galois Field of order , . Nevertheless, also non binary codes can be adopted. Working at upper layer each symbol composing the UL codeword can be made up of packets of bits, depending on the application level.
In order to build the UL-FEC technique on solid ground, the design and analysis has been carried out starting from the Multi Protocol Encapsulation Forward Error Correction Technique (MPE-FEC) adopted by the DVB-H standard , and successively enhanced and modified in the framework of the DVB-SH  standardization group. With respect to the MPE-FEC approach, the implementation of the UL-FEC technique for this framework has required to adapt the parameter setting to the LTE physical layer configurations. In the following, we adopt this terminology:
(i) : the UL block length, that is the number of systematic symbols to be encoded by the UL encoder
(ii) : the UL codeword length, that is the number of UL symbols produced by the UL encoder
(iii) : the actual UL-FEC block length if zero-padding is applied
(iv) : the actual UL-FEC codeword length if zero-padding and/or puncturing is applied
(v) : number of jointly coded channels at physical layer
(vi) : size of each channel in bytes
(vii) : size of the upper layer Cyclic Redundancy Check (CRC) in bytes
(viii) : size of the physical layer CRC in bytes
(ix) : physical layer block length in bytes.
As in MPE-FEC, we define the UL-FEC matrix as a matrix composed of a variable number of rows (n _of_rows) and columns. Each entry of the matrix is an UL-symbol, that is, 1 byte. The first columns represent the systematic part of the matrix and are filled with the systematic UL-symbols coming from the higher level. The last columns carry the redundancy data computed on the first columns. It is worthwhile to notice that the and values depend on the selected UL code rate only, while n _of_rows is a parameter chosen accordingly to the physical layer configuration and is set by using the following formula: . As a consequence, the number of bytes available for each channel in a given UL-FEC matrix column is . With this configuration, the following operations must be sequentially performed.
(1)The information data coming from higher layer are written columns-wise in the systematic data part of the UL-FEC matrix.
(2)A Reed-Solomon (RS) encoding ( ) is performed on each row producing the redundancy part of the UL-FEC matrix.
(3)The data are transmitted column-wise.
(4)An UL-CRC is appended after each group of bytes.
(5)Each group of bytes composes a physical layer information packet.
(6)The PHY-CRC is appended to each physical layer information packet according to the LTE specifications .
For sake of simplicity, we adopt the same RS mother code provided in , which is an RS(255,191). The code rate of this mother code is 3/4. Further code rates can be achieved by using padding or puncturing techniques. For instance, if a UL-FEC rate 1/2 is needed, zero-padding is performed over the last 127 columns of the systematic data part of the UL-FEC matrix, yielding to and . The choice of this RS code allows fully compatibility with DVB-H networks.
It is important to note how the application of the CRC at UL and physical layer has an impact on the overall system performance. To better evaluate this impact, we distinguish to study cases:
(i)Case-A: only the PHY-CRC is considered ( ). In this scenario, the receiver is not able to check the integrity of a single UL packet carried within the same physical layer information packets. This basically means that if error is detected in the physical layer information packet, all UL packets will be discarded;
(ii)Case-B: both PHY and UL CRC are applied.
It is quite obvious that Case-B outperforms Case-A. In fact, if only a small fraction of bits are wrong after physical layer decoding, Case-B is able to discard only the UL packets in which erroneous bits are present, while Case-A discards all carried within the physical layer information packets. The price to pay is an increased overhead of Case-B with respect to Case-A due to the extra CRC bits appended.
At the receiver side, depending whether Case-A or Case-B is taken into account, CRC integrity must be performed at different levels. If the Case-A is considered, only the CRC at physical layer determines the data reliability; whereas in the Case-B, the PHY-CRC could be ignored and the data reliability is only determined by the UL-CRC. Then, the UL-FEC matrix is filled with the reliable data. In particular, for the Case-A an entire column is marked as reliable or not reliable, while in the Case-B the UL-FEC matrix columns could be partially reliable. Finally, the RS( ) decoding is performed on each row. If the number of reliable position in a row is at least , the decoder is able to successfully decode the received information, and all unreliable positions are recovered.
The UL-FEC protection capability against burst of errors can be characterized by the so-called Maximum Tolerable Burst Length (MTBL) , which consists in the maximum time protection that the UL-FEC technique can provide. The MTBL depends on both UL-FEC parameters and PHY data rate. In our proposal one PHY information packet is mapped in one column of the UL-FEC matrix. Since we are dealing with MDS codes, the decoder is able to successfully decode if at least columns are correctly received in the UL-FEC matrix. Thus, the MTBL is simply given by the time taken by transmitting columns, that is, the duration of information packets. The MTBL can be increased by adopting a sliding encoding mechanism . The sliding encoding is a UL interleaver mechanism: a UL-FEC encoder implementing sliding encoding selects the data columns from a window (SW) among the UL-FEC matrices and spreads the parity sections over the same window. Basically, the same effect could be obtained by first normally encoding SW frames and then interleaving sections among the encoded SW frames. The total protection time achievable at upper layer by means of such a technique is given by .
Here, we discuss separately the numerical results obtained by implementing the solutions presented in Section 3. The following general assumptions have been considered during the implementation of all techniques.
The LTE transmitted signal occupies 5 MHz of bandwidth, , located in S-band (central frequency GHz), the sub-carrier spacing is kHz, and FFT/IFFT size is fixed to . The long cyclic prefix is assumed, , thus OFDM symbols are transmitted in each TTI. The resulting OFDM symbols duration is , including the cyclic prefix duration of .
5.1. Inter-TTI Improvements
For evaluating the inter-TTI proposal, the turbo encoder is fed with 2496 information bits, while the circular buffer size is assumed to be 6300, thus resulting in an actual system code rate equal to . All simulations have considered QPSK modulation.
5.2. ACE Performance
The results of the ACE algorithm for PAPR reduction are discussed. First of all, the CCDF of PAPR distribution have been analyzed for verifying the effectiveness of the selected method.
5.3. Redundancy Split Analysis
5.4. End-to-End Performance Evaluation
In this section, the results obtained considering end-to-end simulations in realistic satellite propagation scenario are analyzed. To this aim, we have adopted the Land Mobile Satellite (LMS) channel model proposed in , which is based on measurement campaigns. This channel model is characterized by a three states Markov model. Each state describes different propagation conditions, that are line of sight, moderate shadowing conditions, and deep shadowing conditions. By suitably setting the Markov chain parameters, several environment can be modeled. In our analysis we have considered an elevation angle of 40 degrees and the following environments: open area [O], Suburban [S], Intermediate tree shadow [ITS], Heavy Tree Shadow [HTS]. Such environments are characterized by long fading events due to the superposition of shadowing effects. It is quite obvious that applying the proposed UL-FEC technique without any interleaver working at UL does not allow to cope with such channel impairments. Indeed, the MTBL achievable by adopting UL-FEC without sliding interleaving ( ) is in the order of hundreds milliseconds. To increase the MTBL we adopt the sliding window encoding technique. As already mentioned, this technique basically consists in applying a block interleaver at UL.
(1)perform AWGN simulations (including NL distortion), to obtain the function BLER versus ;
(2)generate the Perez Fontan channel coefficients, obtaining signal levels relative to LOS component;
(3)calculate the received value in LOS conditions;
(4)map the instantaneous value into ;
(5)generate the time series, producing a "1'' (wrong block) or a "0'' (correct block) according to the following algorithm: if [uniform-random-variable BLER ] then time-series-value = 1, else time-series-value = 0.
Adopted LTE Physical layer configurations.
Number of jointly coded channels/number of channel groups
Info-bits per packet
Allocated data carriers per sub-frame [RBs OFDM symbols]
FEC Code rate
Overall Bit Rate Channel
In Figure 12, each curve represents the performance of the QPSK constellation in a given scenario and for a given UL-FEC coding rate. The connected markers in each curve represent the corresponding PHY configurations in a given scenario and for a given UL-FEC coding rate. Regarding the UL parameters, two configuration have been taken into account: rate 1/2 ( , ) and rate 3/4 ( , ). The adopted sliding window size has been set to for the rate 1/2 case, and for the rate 3/4, yielding to a total protection time at UL equal to s, and s, respectively. Notably, for the 16QAM constellation, only one PHY FEC scheme has been considered. Interestingly, the lower UL-FEC protection, that is, 3/4, always outperforms, at the same total spectral efficiency, the higher UL-FEC protection, with the only exception of the Heavy Tree Shadow scenario. In that case, the extremely challenging propagation conditions calls in fact for a very strong protection along with a quite demanding link budget.
The adoption of the 3GPP LTE air interface to broadband satellite networks has been evaluated. The rationale for this choice was the maximization of the commonalities with the terrestrial air interfaces, so as to reduce both non-recurrent engineering and production costs, while easing interworking procedures. The selected numerologies for forward and reverse links are standard compatible. In this sense, the results produced are significant from the 3GPP point of view.
Regarding time domain fade mitigation techniques, one of the major findings consists in a way to obtain the above diversity in an almost standard compatible way. This is the inter-TTI technique, which has been shown to bring significant benefits without touching the physical layer definition.
PAPR reduction algorithms, coupled to predistortion techniques, are a novelty for OFDM transmission through a satellite. We have explored this architecture and our results show that the PAPR itself can be reduced by 2 to 4 dB (guaranteed at 99.9%), which translates into the possibility to reduce the OBO by about 0.7 dB and to gain about 0.5 dB in for typical quality of services. All in all, we can expect a gain in total degradation around 1 dB, which is certainly not negligible.
Regarding frame acquisition procedures, they are quite specific for LTE air interface. The design of acquisition sequences for 3GPP LTE has been performed adapting it to the different requirements set by satellite transmission involving the use of large geographic beams.
Additionally, in order to further extend the link reliability over the satellite link, the use of UL-FEC techniques has been investigated. Simulation results clearly show that the UL-FEC technique is a very effective solution that can drastically improve the achievable block error rate and ESR5(20) performance.
In order to provide useful guidelines for the system design, the analysis of the optimum redundancy split between physical and upper layer coding has been performed. In this case, results show that in most cases it is beneficial to limit the protection at physical layer in order to ease channel estimation and to compensate the reduced performance through a stronger UL coding. The rationale behind this conclusion is that the UL-FEC benefits a larger time diversity thus performing significantly better than the physical layer coding in almost all scenarios.
This work is supported in part by the ESA contract no. 20194/06/NL/US, "Study of Satellite Role in 4G Mobile Networks."
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